The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.
[Patent Document 1]
Japanese Patent Laid-open No. 2003-235259
The present applicant has previously proposed various power supply circuits having a resonant converter on a primary side.
FIG. 16 is a circuit diagram showing an example of a switching power supply circuit having a resonant converter which circuit is formed on the basis of an invention previously devised by the present applicant.
The switching converter of the power supply circuit shown in FIG. 16 is formed by combining an externally excited current resonant converter of a half-bridge coupling system with a partial voltage resonant circuit performing voltage resonant operation only at the time of turn-off in switching.
Two secondary side direct-current output voltages having the same positive and negative levels are output, as shown in the figure.
The power supply circuit deals with load conditions of load power Po=0 W to 150 W, secondary side direct-current output voltage Eo=±35 V or lower, and load current=10 A or lower.
In the power supply circuit shown in FIG. 16, a common mode noise filter including two filter capacitors CL and CL and one common mode choke coil CMC is connected to a commercial alternating-current power supply AC.
As a rectifying and smoothing circuit for generating a direct-current input voltage from the commercial alternating-current power supply AC, a full-wave rectifier circuit including a bridge rectifier circuit Di and a smoothing capacitor Ci is provided in a stage subsequent to the common mode noise filter.
The smoothing capacitor Ci is charged with the rectification output of the bridge rectifier circuit Di, whereby a rectified and smoothed voltage Ei (direct-current input voltage) having a level corresponding to once an alternating input voltage VAC is obtained across the smoothing capacitor Ci.
A switching circuit system formed by connecting two switching devices Q1 and Q2 formed by a MOS-FET by half-bridge coupling as shown in the figure is provided as the current resonant converter supplied with the direct-current input voltage and switching the direct-current input voltage. Damper diodes DD1 and DD2 formed by body diodes are connected in parallel with the switching devices Q1 and Q2 between the drain and the source of the switching devices Q1 and Q2, respectively, in a direction shown in the figure.
A partial resonant capacitor Cp is connected in parallel with the drain and the source of the switching device Q2. The capacitance of the partial resonant capacitor Cp and the leakage inductance L1 of a primary winding N1 form a parallel resonant circuit (a partial voltage resonant circuit) A partial voltage resonant operation, in which voltage resonance occurs only when the switching devices Q1 and Q2 are turned off, is thereby obtained by the partial voltage resonant circuit.
The power supply circuit is provided with an oscillation and drive circuit 2 formed by a general-purpose IC, for example, to switching-drive the switching devices Q1 and Q2. The oscillation and drive circuit 2 has an oscillating circuit and a driving circuit. A drive signal (gate voltage) of a required frequency is applied to the gates of the switching devices Q1 and Q2. Thus the switching devices Q1 and Q2 perform switching operation so as to be turned on/off alternately at the required switching frequency.
An isolated converter transformer PIT (Power Isolation Transformer) transmits the switching output of the switching devices Q1 and Q2 to a secondary side.
One end of the primary winding N1 of the isolated converter transformer PIT in this case is connected to a point of connection (a switching output point) between the source of the switching device Q1 and the drain of the switching device Q2 via a primary side series resonant capacitor C1. Thereby the switching output is obtained.
Another end of the primary winding N1 is connected to a primary side ground, as shown in the figure.
In this case, the primary winding N1 and the series resonant capacitor C1 are connected in series with each other. The capacitance of the series resonant capacitor C1 and the leakage inductance L1 of the primary winding N1 (series resonant winding) of the isolated converter transformer PIT form a primary side series resonant circuit for converting the operation of the switching converter into a current resonance type operation.
According to the description thus far, the primary side switching converter shown in this figure obtains the current resonance type operation by the primary side series resonant circuit (L1-C1) and the partial voltage resonant operation by the partial voltage resonant circuit (Cp//L1) described above.
That is, the power supply circuit shown in this figure employs a form in which the resonant circuit for making the primary side switching converter a resonant converter is combined with another resonant circuit. Such a switching converter will herein be referred to as a complex resonant converter.
Though not described with reference to a drawing, the structure of the isolated converter transformer PIT described above has an EE type core formed by combining E-type cores of ferrite material, for example, with each other. A primary side winding part and a secondary winding part are divided from each other, and the primary winding N1 and a secondary winding N2 are wound around the inner magnetic leg of the EE type core.
A gap having a length of 1.0 mm or less is formed in the inner magnetic leg of the EE type core of the isolated converter transformer PIT to obtain a coupling coefficient of 0.85 or higher between the primary winding N1 and the secondary winding N2.
In practice, the gap G=1.0 mm, and as for the number of turns of the primary winding N1 and the secondary winding N2, the primary winding N1=37 T (turns) and the secondary winding N2=12 T (6 T+6 T with a center tap as a boundary), whereby a coupling coefficient k=about 0.85 is obtained.
An output from the primary winding N1 is induced in the secondary winding N2 of the isolated converter transformer PIT.
The secondary winding N2 in this case is provided with a center tap connected to a secondary side ground as shown in the figure to be divided into a secondary winding part N2A and a secondary winding part N2B.
The secondary winding part N2A is a winding part on the side of a winding termination end part of the secondary winding N2 as a whole. The secondary winding part N2B is a winding part on the side of a winding start end part of the secondary winding N2 as a whole. That is, on the secondary side in this case, the secondary winding part N2B is wound around the central magnetic leg of the isolated converter transformer PIT, and the secondary winding part N2A is wound around the outside of the secondary winding part N2B. In other words, the secondary winding part N2B is wound in a lower part, and the secondary winding part N2A is wound in an upper part.
The secondary winding N2 is connected with two double-wave rectifier circuits formed by a set of a rectifier diode Do1A, a rectifier diode Do2A, and a smoothing capacitor CoA and a set of a rectifier diode Do1B, a rectifier diode Do2B, and a smoothing capacitor CoB shown in the figure.
In this case, of the two secondary side direct-current output voltages Eo having the same positive and negative levels mentioned above, the secondary side direct-current output voltage +Eo of positive polarity is generated by the double-wave rectifier circuit formed by the set of the rectifier diode Do1A, the rectifier diode Do2A, and the smoothing capacitor CoA. The secondary side direct-current output voltage −Eo of negative polarity is generated by the double-wave rectifier circuit formed by the set of the rectifier diode Do1B, the rectifier diode Do2B, and the smoothing capacitor CoB.
The rectifier diode Do1A has an anode connected to the winding termination end part of the secondary winding part N2A, and a cathode connected to the positive electrode terminal of the smoothing capacitor CoA. The rectifier diode Do2A has an anode connected to the winding start end part of the secondary winding part N2B, and a cathode connected to a point of connection between the cathode of the rectifier diode Do1A and the positive electrode terminal of the smoothing capacitor CoA.
The rectifier diode Do1B has a cathode side connected to the winding start end part of the secondary winding part N2B, and an anode side connected to the negative electrode terminal of the smoothing capacitor CoB. The rectifier diode Do2B has a cathode side connected to the winding termination end part of the secondary winding part N2A, and an anode side connected to a point of connection between the rectifier diode Do1B and the negative electrode terminal of the smoothing capacitor CoB.
The negative electrode terminal of the smoothing capacitor CoA and the positive electrode terminal of the smoothing capacitor CoB are connected to each other, and a point of connection between the negative electrode terminal of the smoothing capacitor CoA and the positive electrode terminal of the smoothing capacitor CoB is connected to the secondary side ground.
In these double-wave rectifier circuits, in one half period of an alternating voltage induced in the secondary winding N2, the rectifier diode Do1A conducts to charge the smoothing capacitor CoA with a rectification current I1 shown in the figure, while the rectifier diode Do1B conducts to charge the smoothing capacitor CoB with a rectification current I2.
In another half period, the rectifier diode Do2A conducts to charge the smoothing capacitor CoA with a rectification current I2, and the rectifier diode Do2B conducts to charge the smoothing capacitor CoB with a rectification current I1.
That is, by such an operation, the smoothing capacitor CoA is charged in each half period. The other smoothing capacitor CoB is also charged in each half period.
Then, as shown in the figure, the secondary side direct-current output voltage +Eo of positive polarity is extracted from the positive electrode terminal of the smoothing capacitor CoA. The secondary side direct-current output voltage −Eo of negative polarity is extracted from the negative electrode terminal of the smoothing capacitor CoB.
The secondary side direct-current output voltage +Eo obtained by the smoothing capacitor CoA and the secondary side direct-current output voltage −Eo obtained by the smoothing capacitor CoB are each supplied to a load side not shown in the figure.
In this case, the secondary side direct-current output voltage +Eo obtained on the smoothing capacitor CoA side branches off to be input also as a detection voltage for constant-voltage control to a control circuit 1.
The control circuit 1 outputs a control signal as a voltage or a current having a level varied in such a manner as to correspond to the level of the secondary side direct-current output voltage +Eo to the oscillation and drive circuit 2.
The oscillation and drive circuit 2 changes the frequency of a switching driving signal applied to the gates of the switching devices Q1 and Q2 by varying the frequency of an oscillating signal generated by the oscillating circuit within the oscillation and drive circuit 2 on the basis of the control signal input from the control circuit 1. Thereby the switching frequency is varied. By thus variably controlling the switching frequency of the switching devices Q1 and Q2 according to the level of the secondary side direct-current output voltage +Eo, the resonant impedance of the primary side series resonant circuit is changed, and energy transmitted from the primary winding N1 forming the primary side series resonant circuit to the secondary side is varied. Hence, the level of the secondary side direct-current output voltage −Eo is consequently controlled variably together with the secondary side direct-current output voltage +Eo. That is, constant-voltage control is performed on both secondary side direct-current output voltage +Eo and the secondary side direct-current output voltage −Eo.
Incidentally, a constant-voltage control system that achieves stabilization by thus variably controlling the switching frequency will hereinafter be referred to as a “switching frequency control system.”
FIG. 17 shows operating waveforms of principal parts of the circuit shown in FIG. 16 as results of an actual experiment on the circuit shown in FIG. 16.
FIG. 17 shows experimental results when load power Po on the load side is set constant at 150 W (maximum load power) and alternating input voltage VAC is set constant at 100 V.
Incidentally, in obtaining the experimental results shown in the figure, parts of the circuit shown in FIG. 16 were selected as follows.
Isolated converter transformer PIT    gap length=1.0 mm, and coupling coefficient k=0.85    Primary winding N1=37 T    Secondary winding N2=12 T=secondary winding part N2A+    secondary winding part N2B=6 T+6 T            Primary side series resonant capacitor C1=0.033 μF        Partial resonant capacitor Cp=330 pF        
In FIG. 17, a voltage V1 is a voltage across the switching device Q2, and indicates on/off timing of the switching device Q2. That is, this voltage V1 indicates the switching period of switching operation on the primary side.
As shown in the figure, the peak level of the voltage V1 is clamped at the level of the rectified and smoothed voltage Ei.
A period during which the voltage V1 is at a zero level is an on period during which the switching device Q2 conducts. During this on period, a switching current IQ2 having a waveform as shown in the figure flows in a switching circuit system including the switching device Q2 and the clamping diode DD2. A period during which the voltage V1 is clamped at the level of the rectified and smoothed voltage Ei is a period during which the switching device Q2 is off and the switching current IQ2 is at a zero level as shown in the figure.
Though not shown in the figure, a voltage across the other switching device Q1 and a switching current flowing through a switching circuit (Q1 and DD1) have waveforms obtained by shifting the phases of the voltage V1 and the switching current IQ2 180°. That is, as described above, the switching device Q1 and the switching device Q2 perform switching operation in timing in which the switching device Q1 and the switching device Q2 are turned on/off alternately.
Though not shown in the figure, a primary side series resonance current Io flowing through the primary side series resonant circuit (C1-N1(L1)) flows having a waveform obtained by combining the switching currents flowing through the switching circuits (Q1 and DD1) and (Q2 and DD2) with each other.
Incidentally, the peak level of the switching current IQ2 in this case is 4.6 Ap, as shown in the figure.
Primary side operation with the above-described waveforms being obtained, an alternating voltage V2 having a waveform as shown in the figure is induced on the secondary winding part N2A side (and on the secondary winding part N2B side). In one half period in which the alternating voltage V2 is of positive polarity, the rectifier diodes Do1A and Do1B each conduct as described above. In a half period in which the alternating voltage V2 is of negative polarity, the rectifier diodes Do2A and Do2B each conduct. Thereby, the rectification current I1 flowing on the secondary winding part N2A side and the rectification current I2 flowing on the secondary winding part N2B side each flow having a waveform as shown in the figure in each positive or negative half period.
Incidentally, in this case, the peak level of the rectification current I1 was 8 Ap. The peak level of the rectification current I2 was 3.1 Ap.